1. The Field of the Invention
The present invention relates to the field of voltage reference circuits. In particular, the present invention relates to circuits and methods for providing a voltage reference that uses a metal-silicon Schottky diode for the Complementary proportional To Absolute Temperature (CTAT) voltage source that is added to a properly amplified PTAT voltage source to form a temperature stable voltage reference for low voltage applications.
2. Background and Related Art
The accuracy of circuits often depends on access to a stable bandgap voltage reference. A bandgap voltage reference is a voltage reference approximately equal to the bandgap potential (VG0) of the semiconductor at zero degrees Kelvin.
The bandgap voltage reference circuit is often configured by adding two voltages together: one that is inversely or Complementary proportional To Absolute Temperature (CTAT), and one that is Proportional To Absolute Temperature (PTAT). The CTAT voltage decreases approximately linearly with absolute temperature, whereas the PTAT voltage increases approximately linearly with absolute temperature.
The CTAT voltage source is typically the base-emitter voltage (VBE) of a diode-connected bipolar transistor. FIG. 5 illustrates a plot of the base-emitter voltage (VBE) represented on the vertical axis as a function of absolute temperature in degrees Kelvin represented on the horizontal axis. The slope of the base-emitter voltage VBE versus temperature is dependent on the current density through the bipolar transistor. For example, approximate line 501 represents the VBE versus temperature function when the current density is relatively low; approximate line 502 represents the VBE versus temperature function when the current density is moderate; and approximate line 503 represents the VBE versus temperature function when the current density is relatively high. In each case, however, the base-emitter voltage (VBE) at zero degrees Kelvin (i.e., the Y-intercept of FIG. 5) is equal to the bandgap of the semiconductor at zero degrees Kelvin (VG0). In the case of FIG. 5, VG0 is shown as 1.2 volts which approximates the bandgap voltage for silicon at zero Kelvin.
A close approximation to this relationship is shown in the following Equation 1:                     VBE        =                  VGO          -                                    V              T                        ⁢            L            ⁢                                                  ⁢                          N              ⁡                              (                                                      I                    O                                                        I                    D                                                  )                                                                        (        1        )            
Where,                ID is the diode current;        IO is a process and geometry specific current approximately twenty orders of magnitude higher than the diode reverse saturation current, IS, for the semiconductor (IO is usually significantly higher than the diode current ID);        VT is the thermal voltage which is equal to kT/q, where k is the well-known Boltzmann constant, T is absolute temperature, and q is the well-known charge of an electron.        
To form a PTAT voltage, the difference between the base-emitter voltage (VBE) of two bipolar transistors is used, where the current density is different for each bipolar transistor. As shown in FIG. 6, the difference between VBE for the high biased diode corresponding to line 503 and the low biased diode 501 is a linearly increasing function with a Y intercept of zero.
The curve depicted in FIG. 6 can be described by the relationship described in Equation 2 as follows:                               Δ          ⁢                                          ⁢                      V            BE                          =                              V            T                    ⁢          L          ⁢                                          ⁢                      N            ⁡                          (                                                J                  2                                                  J                  1                                            )                                                          (        2        )                            where J1 and J2 are the respective current densities flowing through the emitters of the transistors, and is equal to the current flowing through the emitter IE divided by the emitter area AE.        
In order to form the bandgap voltage reference, the PTAT voltage (VPTAT, in this case ΔVBE) is multiplied by a constant G. The result is added to the CTAT voltage (VCTAT, in this case VBE) to obtain the output voltage VOUT. This is represented mathematically by the following Equation 3:VOUT=VCTAT+G·VPTAT (3)and also by the following Equation 4:VOUT=VBE(J1)+G·[VBE(J1)−VBE(J2)] (4)
The constant G is chosen to make the slope of G•VPTAT versus temperature equal in magnitude but opposite in sign to the slope of VCTAT versus temperature. This yields a voltage VOUT which is substantially independent of temperature as depicted in FIG. 7, and is approximately equal to the bandgap potential of the semiconductor.
FIG. 8 schematically illustrates a conventional circuit 800 that produces the relationship described by Equation 4. This conventional circuit 800 is especially employed in Silicon CMOS processes in which parasitic PNP bipolar transistors are available having a substrate that serves as the collector. The conventional circuit 800 includes two bipolar transistors 801 and 802.
The current density J1 passing through bipolar transistor 801 is equal to the current I1 divided by its emitter area A1. The current density J2 passing through bipolar transistor 802 is equal to the current I2 divided by its emitter area A2. The voltage at the emitter terminal of bipolar transistor 801 (i.e., VBE(J1)) is provided to the positive input terminal of the amplifier 803. The voltage at the emitter terminal of bipolar transistor 802 (i.e., VBE(J2)) is provided to the negative input terminal of the amplifier 803. The amplifier 803 has gain G. Accordingly, a voltage of G•(VBE(J1)−VBE(J2)) is applied at the output terminal of the amplifier 803. The output voltage VOUT is obtained by summing 807 the output voltage of the amplifier 803 with the base-emitter voltage of the bipolar transistor 801.
The currents I1 and I2, and the emitter areas A1 and A2 are chosen such that the voltage VBE(J1) at the emitter terminal of bipolar transistor 801 is larger than the emitter voltage VBE(J2) at the emitter terminal of bipolar transistor 802 and such that the difference in base emitter voltages (i.e., VBE(J1) minus VBE(J2)) is significantly larger than the offset voltage of the amplifier 803.
The current sources 805 and 806 used to bias the respective bipolar transistors 801 and 802 are typically generated using the output voltage VOUT of the bandgap reference circuit 800. If the supply voltage does not affect the currents through either bipolar transistor, the output voltage is independent of the supply voltage as well as temperature for higher supply voltages.
In order to minimize the dependence of the output voltage VOUT on temperature, the bipolar transistors 801 and 802 should be carefully matched. Matching of devices is quite difficult. Minor and yet inevitable spatial process variations often cause some mismatch between common devices.
FIG. 9 schematically illustrates an alternative bandgap voltage reference circuit 900 along with an associated timing diagram 910. The bandgap voltage reference circuit 900 only uses one bipolar transistor 901. Accordingly, there is no matching issue between two bipolar transistors as there is with the bandgap voltage reference circuit 800 of FIG. 8. Furthermore, power consumption is reduced since there is only one bipolar transistor drawing current. The bandgap voltage reference 900 requires a low frequency clock signal φ1, and a non-overlapping complement, φ2.
During the period when φ1 is high, a higher current I1 is passed through the bipolar transistor 901 creating a higher base-emitter voltage VBE(J1) which is sampled and stored on capacitor C1. J1 is the current density through the bipolar transistor 901 when the total current is I1. During the period when φ2 is high, a lower current I2 is placed through the bipolar transistor 902 generating a lower base-emitter voltage VBE(J2) which is sampled and stored on capacitor C2. Again, to generate the relationship described by Equation 4, the difference between these two voltages is multiplied by a specific gain G using an amplifier 903. The amplified voltage is then added to the higher VBE voltage. Once again, the amplifier gain G is chosen such that the resulting output voltage VOUT is a constant with respect to temperature.
Each of these conventional bandgap voltage reference circuits 800 and 900 are effective in generating a bandgap voltage reference that is approximately equal to the bandgap potential of the underlying semiconductor as long as the high supply voltage is sufficiently high for the amplifiers 803 and 903 to generate voltages below and approaching the bandgap potential (1.2 volts in the case of silicon). Accordingly, as supply voltages drop to and below 1.2 volts, the performance of circuits 800 and 900 will degrade. With lower voltage applications becoming more prevalent, voltage references that are lower than the bandgap potential of the semiconductor may be useful.
Accordingly, what would be advantageous are silicon-based voltage reference circuits that provide voltage references that are relatively independent of temperature and below the bandgap potential of silicon. It would especially be advantageous if such reference circuits may operate with lower supply voltages.